Self-adjusting gate bias network for field effect transistors

ABSTRACT

The present invention is directed to a self-adjusting gate bias network for field effect transistors in radio frequency applications. A bias network for field effect transistors is provided comprising a field effect transistor having a source electrode connected to ground and a drain electrode connected to a load; a radio frequency network connected to the gate electrode; a gate bias network connected to the gate electrode; wherein a device having a non-linear characteristic is provided in series between the gate electrode and the gate bias network such that a forward bias current at the gate electrode of the field effect transistor is reduced or prevented. 
     The reduction or prevention of a forward bias current leads in overdrive conditions to a self-adjustment of the bias point of the field-effect transistor improving the reduction of distortions of an amplifier or changing the class of oscillators connected to the gate electrode.

The present invention is directed to a self-adjusting gate bias networkfor field effect transistors, specifically a self-adjusting gate biasnetwork for field effect transistors in radio frequency applications.

BACKGROUND OF THE INVENTION

In radio frequency (RF) applications of field effect transistors, inaddition to a radio frequency signal, a DC network is connected to thegate and the drain electrode of a transistor providing a gate and adrain bias to set the operating point of the transistor.

A typical example of a field effect transistor with a gate bias networkis shown in FIG. 1 a and FIG. 2 a. A gate electrode of a field effecttransistor 1 is connected to a radio frequency source 3 and a DC or lowfrequency network 4. The DC network 4, here consisting of a DC source,sets the operating point of the transistor. As shown in FIG. 1, theradio frequency network 3 and the DC network 4 are usually coupled tothe gate electrode of the field effect transistor via a bias-T device 6.The bias-T 6 feeds the DC to the transistor 1 and also decouples the RFinput and output path. The bias-T usually consists of an RF choke(inductor) or RF block 13 in series at the DC path which acts as athrough for DC, and a capacitor or DC block 14 in series at the RF pathwhich blocks DC current to flow towards the RF source 3 but at the sametime lets the RF pass through to the gate electrode of the transistor 1.The source of the transistor 1 is connected to ground. The transistordrain is also coupled via a drain bias-T 7 to a drain DC network 8, herejust a DC source, and a load 2.

Under normal operating conditions, the transistor 1 the DC voltage atthe gate is kept negative, for a High Electron Mobility Transistor(HEMT) usually between −5 to 0 V. A HEMT is a field effect transistorincorporating a junction between two materials with different band gaps,i.e. a heterojunction, as the channel. The amount of negative voltage ofthe gate bias network 4 controls the current flow in the channel to thedrain. The negative voltage varies for different FET according to designand physical properties. Due to the negative voltage, a very smallnegative current flows from the DC network 4 to the gate electrode oftransistor 1, indicated by the full black arrow in FIG. 2 a. FIG. 2 a isa schematic diagram of the same network as in FIG. 1 without the detailsof the bias-T's 6 and 7. A so called pinch-off condition of the DCnetwork 4 is defined by the negative voltage when the channel is totallyoff, i.e. no DC current flows into the channel or to the load connectedto the drain. An example of a radio frequency network 3 in FIG. 1 may bea power amplifier or an oscillator feedback circuit for establishingoscillations.

In power amplifier applications, the linearity specification is veryimportant to prevent neighbouring channel interference. Linearity is,e.g., a crucial issue in modern wireless communication. Under normaldriving conditions of the FET 1 in FIG. 1 and FIG. 2 a, the RF signal 3at the input is very small and the device 1 operates in its linearregion, thus signal distortion is very small. In such a case, a verysmall negative current flow towards the gate of the device 1 and a highpositive drain current flow towards the drain, see FIG. 2 a.

However, in communication front ends under overdrive conditions, thedistortion components may disturb the wireless communication in a numberof different ways. The amplifier may be overdriven by a very large inputsignal. As a consequence, the amplifier may produce distortioncomponents, which interfere its neighbouring channel. The overdrivinginput signal may also result from an undesired origin. Possibleundesired signals could be a large jamming signal from radar or anyother transmitter. The large input signal may also come from thetransmitter amplifier in a transceiver system. In FIG. 2 b an overdrivecondition of the device of FIG. 2 a is shown, where the gate current ofthe FET 1 increases and becomes eventually positive as indicated by thearrow in the gate DC path. In this condition, the device 1 produces veryhigh distortion products, which may damage the device.

In such overdrive applications, it would be desirable to adjust theoperating point of the transistor or the DC source voltage to suppressdistortion components.

Another very important application of field effect transistors inRF-engineering are oscillators. Here, not a signal from a poweramplifier, but a signal from a feedback circuit may be connected to thegate electrode of a field effect transistor to generate periodicalsignals, the main task of an oscillator circuit. In most cases a sine intime domain is provided at the oscillators output. Square or triangleare also familiar wave shapes. Oscillators are available for a number ofvery different applications having very different properties. Incommunication applications, an oscillator with a very low phase noise isimportant. In case of large power generation, the noise in phase is lessimportant.

In order to start and/or maintain the oscillation, a certainamplification of the signal is required. Therefore, the transistors biaspoint often destines the class of operation. Basically, a transistor maybe operated in different classes. Several classes are known: A, AB, B,C, D, E, F. In class A, the linearity is very good, however theefficiency is very bad. It may theoretically up to 50%, but in mostapplications the class A efficiency is less than the ideal value of 50%.With classes AB, B, C, E, F the efficiency increases but the linearitydecreases. Different classes of operation have different properties. Foroscillator design, class A is very nice and easy to realize. In class C,the design becomes more difficult, because the start up condition is notfulfilled.

Therefore, also in oscillator applications of a field effecttransistors, the setting of the bias point is important and adjustmentin order to establish and maintain oscillations desirable.

SUMMARY OF THE INVENTION

It is an object of the present invention to provide a self-bias networkfor field effect transistors. It is a further object of the invention toincrease the robustness of a field effect transistor in relation tooverdrive situations of an amplifier or an oscillator. A further objectof the invention is to provide a bias network for field effecttransistors, which changes the class of an oscillator during start-up ofthe oscillation by itself.

Accordingly, a bias network for field effect transistors is provided,comprising a field effect transistor, the field effect transistor havingan internal gate diode, a source electrode connected to ground and adrain electrode connected to a load; a radio frequency network connectedto the gate electrode of the field effect transistor; a gate biasnetwork connected to the gate electrode of the field effect transistor;and a device having a non-linear characteristic is provided in seriesbetween the gate electrode of the field effect transistor and the gatebias network. The radio frequency network and the gate bias network areconnected to the gate electrode of the field effect transistor via agate bias-T device and wherein the device having non-linearcharacteristic is provided between the gate bias-T device and the gatebias network.

Advantageously, such use of a device having non-linear characteristicsleads to a self-adjustment of the bias point of the field effecttransistor. In the case that the RF network is an amplifier, this leadsan improved linearity of the field effect transistor in overdrivesituations. Using the proposed bias, the power amplifier may berestricted from being over distorted.

The device having a non-linear characteristic may be provided such thata forward bias current at the gate electrode of the field effecttransistor is inhibited or reduced.

In case the RF network is an oscillator feedback circuit, afterswitch-on of the oscillator the self-adjustment of the bias point leadsto a change of the oscillator class.

Preferably, the gate bias network provides a negative DC voltage to thegate electrode of the field effect transistor and the device having anon-linear characteristic is arranged such that a negative currentpasses through the device but a positive current is blocked.

The device having a non-linear characteristic may further comprises adiode or a transistor or a network having the characteristic of a diode.The field effect transistor may be a HEMT transistor.

A second bias-T device may be provided between the drain electrode andthe load as well as between the drain electrode and a drain biasnetwork.

The radio frequency network may input the output signal of a poweramplifier to the field effect transistor. The radio frequency networkmay instead input the output signal of a feedback network such that anoscillator network is established, wherein the self-adjustment of thebias network leads to a change of class of the oscillator afterstart-on.

The device having non-linear characteristics may be realized as a diode,wherein a zener diode may be connected to a point between the diode andthe bias-T such that the DC voltage is limited in the negativedirection.

Accordingly, a method for self-adjustment of a bias point of a fieldeffect transistor in RF applications, comprising the steps of: providinga negative DC voltage by a gate bias network to a gate electrode of afield effect transistor and providing an additional RF signal by a RFnetwork to the gate electrode such that a negative current flows towardsthe gate of the field effect transistor; Increasing the RF signal of theRF network, whereupon the gate current increases to become positive;Self-adjusting the bias point of the field-effect transistor by blockingthe gate current by the device having non-linear characteristics fromreaching the gate electrode, whereupon the DC voltage at the gateelectrode of the field effect transistor becomes more negative.

An oscillator feedback-circuit may be connected to the gate electrode ofthe transistor and the self-adjustment of the bias point may lead to achange of the oscillators class.

The method may further comprise the step of limiting the maximumnegative value of the self-adjustment of the bias point by providing azener diode between the device with non-linear characteristics and thegate electrode.

In addition, the use of a device having non-linear characteristics in abias network of a field effect transistor is claimed for theself-adjustment of the bias point of the field effect transistor,wherein the device is provided in series between a gate electrode of thefield effect transistor and a gate bias network such that a forward biascurrent at the gate electrode of the field effect transistor isinhibited.

BRIEF DESCRIPTION OF THE DRAWINGS

For a better understanding of these and other objects of the presentinvention, reference is made to the detailed description of theinvention, by way of example, which is to be read in conjunction withthe following drawings, wherein like elements are given like referencenumbers, and wherein:

FIG. 1: shows an example of a typical bias network for a field effecttransistor with details of drain and gate bias-T devices in the gate anddrain path of the field effect transistor, respectively;

FIG. 2: shows the same example as in FIG. 1 but without the details ofbias-T devices in the gate and drain path under (a) normal operatingconditions and (b) overdrive conditions;

FIG. 3: shows a first embodiment of the present invention under (a)normal operating conditions and (b) overdrive conditions;

FIG. 4: shows the output spectrum of a device excited with a two-tonesignal;

FIG. 5: shows details of a preferred embodiment of the inventive gatebias network inhibiting a forward bias current at the gate electrode ofthe field effect transistor;

FIG. 6: shows details of an alternative embodiment of the inventive gatebias network of the present invention;

FIG. 7: shows the C/I ratio over input driving power of the firstembodiment of the present invention;

FIG. 8: shows the C/I ratio over output power of the first embodiment ofthe present invention;

FIG. 9: shows the output power and IM3 power of the first embodiment ofthe present invention (other data same as in FIG. 8);

FIG. 10: shows a second embodiment of the present invention, anoscillator circuit using the inventive bias network;

FIG. 11: shows a changing of the bias point during oscillation switch onfor the second embodiment of the invention of FIG. 10;

FIG. 12: shows the gate current vs. gate-source voltage of transistor 1in FIGS. 3 and 10;

FIG. 13 shows in the left part FIG. 12 rotated by 90°, and in the rightpart the gate-source voltage versus time (lower right part) as well asthe gate current vs. time (upper right part) of transistor 1;

FIG. 14: shows the gate-source voltage vs. time of transistor 1;

FIG. 15: shows the gate-source voltage vs. time (lower part) and thegate voltage vs. time (upper part).

DETAILED DESCRIPTION OF THE DRAWINGS

In the following, reference is made to a HEMT transistor as a fieldeffect transistor. However, it is pointed out that such reference isonly made by way of example and any field effect transistor may beemployed throughout the different embodiments of the present invention.

FIG. 3 shows a first embodiment of the present invention under (a)normal operating conditions and (b) overdrive conditions.

The gate electrode 1 a of a field effect transistor 1 is connected to agate DC bias network 4 and a radio frequency network 3, wherein thesource electrode 1 c is connected to ground and the drain electrode 1 bis connected to a drain DC bias network 8 and a load 2.

The gate and drain paths usually comprise a gate bias-T 6 and a drainbias-T 7 for coupling the RF signal 3 and the DC signal 4, but thisinvention is not restricted to the use of bias-Ts. Instead, any othercomponent, which allows the coupling of RF and DC signals while lettingthrough the DC signal and the RF signal to the gate electrode 1 a butpreventing the DC component to reach the radio frequency network 3 maybe used.

The inventive DC bias circuit is similar to a commercial bias network ofFIG. 1, but between the gate bias network 4 and the gate electrode 1 a,a device 5 having non-linear characteristics is provided such that aforward bias current at the gate electrode 1 a of the field effecttransistor 1 is inhibited. The field effect transistor 1 is preferablyan AlGaN/GaN HEMT. At normal operating conditions, the gate 1 a of FET 1is kept at a negative voltage at a desired bias condition. At thiscondition, a very small negative current flows towards the gate 1 a,i.e. means the internal gate-diode of the FET 1 is in reverse bias. Thenegative voltage of the DC network 4 at gate electrode 1 a delivers anegative current to the gate electrode 1 a. It should be noted that theinset graph of the gate voltage Vg in FIG. 3 is only schematically anddoes not show any sign of the voltage, which is usually negative.

It was surprisingly found that with increasing RF drive signal from theRF network 3 at the gate electrode 1 a, the FET 1 enters to a so-calledself-bias condition and the previously negative current startsincreasing towards zero. When RF drive 3 at the gate 1 a becomes veryhigh, the gate current even tries to shift towards positive values. Inthis operating condition, the internal gate-diode of the FET 1 operatesin forward bias. The FET 1 cannot stand this condition and is easilydamaged.

The invention restricts this current by using a diode 5. As aconsequence, in order to fulfil the diodes 5 zero current condition, theFET 1 is automatically pinched off. As a surprising result, this meansthat the DC voltage at the gate 1 a of the device becomes even morenegative, leading to a self-adjustment of the gate bias point of the FET1.

It is important to notice that the “gate-diode” mentioned above refersto the Schottky diode at the gate 1 a of the FET 1, which is an internaldiode of the FET 1 and forms a part of the device 1 itself. The device 5or DC path diode refers to an external p-n diode that is introduced intothe DC path 4 as a part of the bias circuitry.

Preferably, the forward bias inhibiting device 5 is realized by a diodearranged such that it lets a negative current pass through to the gateelectrode 1 a of the field effect transistor 1 but blocks a positivecurrent from reaching the gate electrode 1 a. However, the preferredreference to a diode is of no restriction to the present invention. Ingeneral, any two- or three-terminal device allowing the prevention of apositive current to reach the gate 1 a of the FET 1 may be employed. Thedevice 5 could be a diode as a two-terminal device as displayed in FIG.5 for the preferred first embodiment. It could also be realized by atransistor as a three terminal device. Also, a sensor, which senses apositive current and digitally regulates the bias voltage may be used.The diode is the simplest device allowing the blocking of a positivegate current, which leads to a self-adjustment of the bias point of theFET 1. Furthermore, additional resistors in series and parallel to thedevice 5 may be used to realize a source impedance characteristicbetween pure voltage control and an ideal diode. FIG. 6 shows analternative embodiment of the device 5. Parallel to the series connecteddiode between the negative gate bias network 4 and the bias-T 6 isarranged a resistor and a zener diode with a resistor in series isconnected to a point between the device 5 and the bias-T 6. The zenerdiode limits the negative voltage in amplitude and controls the biaspoint. In case of very high overdrive signals, the negative voltage atthe gate of the device can become extremely high. Using the Zener diodeand the resistance between gate and ground, this negative voltage can beset to a certain limit according to the breakdown voltage of the zenerdiode and the resistance. The resistance between the gate and grounddissipates the current from gate and prevents a further negativecharging of the gate. The resistors provide, in cooperation with thecapacitor of the gate bias-T 6, a bandwidth to define the speed of thereaction. The bias T is primarily a low pass filter. The resistance andthe capacitance value define the RC time constant and thus the reactiontime for the pinch off. For instance, if the initial bias is −2V andchanges to −6V at overdrive conditions, the time for the voltage to gofrom −2V to −6V depends on the RC time constant and may take a fewhundred nanoseconds This may be important for amplifiers in AM operationmode or other fast variations in the level of drive. Also, RLC networksrealizing the desired transient behaviour may be used. Thus, throughoutthe following, any reference made to a diode may be replaced by theabove-mentioned devices and networks.

The self-adjustment of the gate bias is described in the following byreference to FIGS. 12-15. The self-adjustment employs the non-linearityof the internal gate-source Schottky diode of the junction field effecttransistor 1 having non-linear input characteristics. A junction-FET 1is a voltage driven component.

The direct current gate-source characteristic line of the FET 1 of FIG.11 is similar to that of a diode, resulting from the general design of afield effect transistor. For junction-FETs, the gate of the transistoris realized as a PN-contact or a metal-semiconductor contact. Bothimplementations of the gate result in the formation of an internaldiode, a PN-diode or Schottky diode.

Now, transistor 1 is usually driven at operating points at which thegate current is negligible small and negative. The gate current Ig isthe result of the transfer function of the internal diode characteristicline. The gate current Ig is composed of a negative direct current partand an alternating current part, the latter shown in the lower rightpart of FIG. 13. For instance, the direct current operating pointvoltage may be −2V, a voltage at which the direct current would benegative, see FIG. 12 or left part of FIG. 13, and the alternatingcurrent part may be 1V*sin(ωt). The resulting current would consequentlyfollow the equation Ugs=−2V+1V*sin(ωt). As can be seen in the upperright part of FIG. 13, the gate current Ig is small and negative.

The function employed in the present application is based on therequirement of an internal, non-linear diode characteristic line oftransistor 1, the characteristic line preferably following anexponential function. If now the alternating current, e.g. a sinusoidalalternating current, at the gate strongly increases, the sum of thedirect current and the alternating current may become positive incertain parts, e.g. the maxima of the alternating current. Consequently,due to the non-linear characteristic of the internal gate-diode oftransistor 1, the shape of the gate current becomes non-sinusoidalhaving larger positive areas A1 than negative areas, as shown in FIG.14. FIG. 14 shows the gate current in the upper part and the sinusoidalgate-source voltage in the lower part. The positive region A1 under thegate current line thus becomes larger than the negative region. Thus,the gate current now changes sign from negative to positive currents.This sign change is employed in the present invention by placing a diode5 between the DC part of the gate bias network 4 and the field effecttransistor 1, which limits the gate current to the negative region. Thisprevention of a positive gate current results in a DC self-adjustment ofthe operating point of transistor 1. Since the positive current is notallowed to pass diode 5, it charges capacitor C1 to more negativevalues, which is equivalent to a change of the gate-source voltage Ugs.This can be seen in FIG. 15 by the thick black line in the lower part,which shows the DC part of the gate-source voltage, i.e. the operatingpoint. The direct current part Ugs changes slowly with respect to theperiod length of the alternating current signal shown. The directcurrent charges capacitor C1 of the bias network with a different andlower voltage until the positive part of the gate current is equal tothe negative part of the gate current, i.e. until no DC part exists.This re-charging of capacitor C1 leads to the self-adjustment anddepends on the height of the amplitude of the input signal. If theDC-part of current Ig is negative, the current is able to flow betweenthe source of the DC-part 4 and transistor 1. If the current ispositive, the DC-part 4 is not able to steadily flow into transistor 1since the current is blocked by the inventive diode 5. The positive DCcurrent of Ig thus comes from capacitor C1. The capacitor isconsequently re-loaded and changes its voltage value, which is identicalwith the operation point of transistor 1.

In the example, the capacitor C1 is charged from −2V to −3V over timeresulting in a shift of the operating point of transistor 1 from −2V to−3V and smaller amplification of the positive part of the gate currentIg as can be seen from the upper part of FIG. 15. Please note that thesize of the positive areas A1 of the gate current Ig decreases overtime.

To conclude, upon a significant increase of the alternating current partat transistor 1 which usually would result in distortions, theemployment of a diode 5 between the gate of transistor 1 and the DC biasnetwork 4 results in a self-adjustment of the operating point oftransistor 1 and thus a prevention of distortions.

A possible application of the RF source network 3 may be an amplifier.As initially stated, linearity is a crucial issue in modern wirelesscommunication. Power amplifier linearity specification is very importantto assure neighboring channel interference. If the amplifier isoverdriven, it generates very high distortion components, which maycompletely block the neighboring channel or desensitize the front endLNA. Using the proposed bias network including the series connecteddevice 5, the power amplifier can be restricted from being overdistorted.

If an RF amplifier is used to amplify a pure sine wave then the outputsignal consists of the amplified signal and higher harmonics. Thenearest harmonic product occurs at the double of the signal frequencyand thus can be filtered out easily. For a modulated signal, however,this does not work anymore because mixing products are generated whichfall in the signal band and it is impossible to filter them out. Thesemixing products are called intermodulation distortion products. Thesimplest signal having such characteristics is a two-tone signal, whichis similar to an amplitude-modulated signal. A two-tone signal consistsof two closely spaced sine waves. The following equations and FIG. 4illustrate the intermodulation distortion mechanism.

If we consider an amplifier or any device that has a nonlinear transfercharacteristic, then the output of the device can be written as thefunction of input as a power series:

V _(out) =a ₀ +a ₁ ·V _(in) =a ₂ ·V _(in) ² +a ₃ ·V _(in) ³ + . . . a_(n) ·V _(in) ^(n) +  (1)

A simple two-tone signal having the same amplitude A can be written as:

f ₁ +f ₂ =A(sin ω₁ t+sin ω₂ t)  (2)

The next step is to replace Vin in equation (1) by the two-tone signalas shown in equation 2 and to do some mathematical manipulations to havea spectral representation. FIG. 4 shows the resulting frequencycomponents considering the terms up to third order in equation 1,Table-1 adds details on the respective spectral lines.

TABLE 1 List of output frequency components of a nonlinear systemexcited by a two-tone signal. The phase of a term is defined by itsbehavior at t = 0. Frequency components Magnitude Phase dc a₀ + a₁ · A². . . f₁, f₂ ${a_{1}A} + {\frac{9}{4}{a_{3} \cdot A^{2}}}$ Sin 2f₁,2f₂ $\frac{1}{2}{a_{2} \cdot A^{2}}$ -Cosine 3f₁, 3f₂$\frac{1}{4}{a_{3} \cdot A^{3}}$ -Sin (f₁ + f₂), (f₂ − f₁) a₂ · A²-Cosine, Cosine (2f₁ + f₂), (2f₂ + f₁) $\frac{3}{4}{a_{3} \cdot A^{3}}$-Sin (2f₁ − f₂) $\frac{3}{4}{a_{3} \cdot A^{3}}$ Sin for 2f₁ 

 f₂ Otherwise −sin (f₁ + 2f₂), (f₁ − 2f₂)$\frac{3}{4}{a_{3} \cdot A^{3}}$ Sin for 2f₂ 

 f₁ Otherwise −sin

The components at (2f₁−f₂), (2f₂−f₁) are of special interest becausethey fall inside the signal band. They are commonly referred to asthird-order intermodulation products (IM3). (2f₁−f₂) is called “IM3L”and (2f₂−f₁) is called “IM3R” where “L” and “R” stand for Left and Rightsideband. If the input signal is increased by a factor x then IM3products will increase by x³. That means in a flat gain operation, ifthe input and output fundamental signal amplitude is increased by 1 dB,the third-order components grow by 3 dB. If we imagine that we couldincrease the input signal amplitude further and further whilemaintaining this slope, at some point the IM3 will meet the Pout curve.This point is defined as third-order intercept point, often abbreviatedas IP3 or TOI. This point is normally a point derived by extrapolation.A real amplifier will reach saturation far below this power level. TheIP3 has been the most widely used measure for defining device linearity.Intercept points of any order can be calculated by a single RFmeasurement when applying the following formulas,

$\begin{matrix}{{{IP}_{n} = {{Pout}_{({fundamental})} + \frac{({Suppression})_{n}}{\left( {n - 1} \right)}}}{{where},{({Suppression})_{n} = {{Pout}_{({fundamental})} - {Pout}_{({IMn})}}}}} & (2)\end{matrix}$

For IP3, one obtains the specific result

$\begin{matrix}{{IP}_{3} = {{Pout}_{({fundamental})} + \frac{{Pout}_{({fundamental})} - {Pout}_{({{IM}\; 3})}}{2}}} & (3)\end{matrix}$

This calculation is valid only if the measurement is performed carefullywithin the linear operation regime ensuring that the distortion productsare well above the noise level.

C/I ratio is carrier to intermodulation ratio which is the differencebetween the power of carrier signal and the distortion signal.

C/I _(—) L=Pout_((Fund) _(—) _(L)) −Pout_((IM3) _(—) _(L)) and

C/I _(—) R=Pout_((Fund) _(—) _(R)) −Pout_((IM3) _(—) _(R))

That means, the larger the value of C/I, the better the linearity, orthe distortion compared to the signal strength is low. This measure ismore realistic than IP3 as it deals with a complete power sweep. This isnot an imaginary extrapolated number. Therefore, it is of specialinterest to see the distortion at overdrive condition in the presentinvention.

By using the proposed bias network, the distortion products aresuppressed by a significant amount while this condition does not hampernormal operation, i.e. when the device operates in the linear region.

In the following figures some measurement examples are provided. In FIG.7, the input RF power is swept from a very small power to very largeoverdrive. The C/I ratio over input driving power of FIG. 7 is measuredat 2 GHz with a tone spacing of 10 MHz, Vd=28V and Vg=−1.8 V (at DCsource). In FIG. 8, the C/I ratio is shown over output power.

As can be seen from FIG. 7, the overdrive region starts at about 12-14dBm input power. After a certain value of Pin, the C/I drops drasticallyin case of operation with normal bias network 4 of FIG. 2 resulting invery high distortion. In case of the inventive bias network 4 withdevice 5, the C/I ratio does not fall anymore at an overdrive conditionabove 13 dBm. The new bias network automatically pinches off the deviceand the negative voltage at the gate 1 a becomes even more negative. TheDC gate voltage of the DC bias network 4 at the device 1 is also plottedin FIG. 7. Under normal conditions, the DC gate voltage is −1.8 V.However, with increasing RF input power from RF network 3, at about 13dBm the DC gate voltage drops and reaches −3 Volt at about 17 dBm. Itcan be seen that the C/I curve inverts its direction in case of theinventive self-adjusting bias network with device 5. This is because theoutput-power slightly drops due to pinching-off. FIG. 9 shows the outputpower and IM3 power (other data same as in FIG. 8). The right scale(IM3) is weighted three times more than the left (Pout) scale. Thecircles indicate which scale applies. However, this device 5 should notbe understood as a replacement of a linearizer. This work does not dealwith usual linearization. In usual linearization applications, specialcircuitry is used to cancel out or suppress the distortion productskeeping the output power high in the region, in which the amplifierusually operates. In our case, the device is a protection and stops theamplifier to be over distortive in overdrive conditions from an unwantedsource and thus does not interfere with other channels. In other words,linearizers are circuits which straighten the transistor characteristic.Usually, a pre amplifier is used having an inverse characteristic withrespect to the transistor. This is not done here. Instead, thecharacteristic is cut in the overdrive region such that distortions areprevented. It just protects the amplifier 3 to be over-distortive. Thedistortion components can be either from its own input signal or can bedue to any over distorted neighboring amplifier or any kind of jammingsignal.

FIG. 10 is directed to a second embodiment of the present invention, anoscillator circuit using the inventive self-adjusting bias networkcomprising a normal bias network 4 and a device 5 inhibiting a positiveDC current to reach the gate electrode 1 a. It contains an activeelement, preferably a pHEMT. The transistors source 1 c is connected toground. The drain 1 b is connected to a drain power supply 8 via aninductor that provides only a flow of DC-current as part of a drainbias-T 7. Furthermore the transistors drain 1 b is connected to theoscillators load 2, for instance a resistor or output port, via acapacitor as another part of the bias-T 7. The drain electrode 1 b isalso connected to a feedback network 10 via the same capacitor. Thecapacitor kills every DC-portion. Only the RF-power from the feedbackcircuit 10 is able to flow through the capacitor. The feedback's outputis connected to the transistors gate electrode 1 a. It provides a smallpart of the oscillators output signal with a phase that is necessary foroperation. The phase of the whole loop is 0 or a multiple of 2π.

The novel self-adjusting gate bias network is very important for thebenefit of the invention. The transistors gate 1 a is also connected tothe DC gate voltage supply 4 via an inductor and a diode as a preferredembodiment of device 5 in series. The invented use of the diode is tosuppress a current with a wrong direction, here a positive direction.Between these both elements a capacitor C1 is parallel connected toground.

The gate bias voltage 4 is usually negative. For this reason, theDC-current Ig,DC is also negative and is able to flow through theinductor and through the diode 5. The alignment of the diode is fornegative current. With a high RF amplitude of the feedback circuit 10,at the transistors gate 1 a the DC-current Ig,DC would like to becomepositive. The diode 5 as used prohibits a positive current from reachingthe gate electrode 1 a. However, a small part of a positive currentflows, not continuously, but rather only limited in time, into thecapacitor C1. The consequence is a variation of the capacitors voltage,which is equal to a variation of the Gate-Source-voltage.

FIG. 11 shows a changing of the field effect transistors bias pointduring oscillation switch-on for the second embodiment of the inventionas shown in FIG. 10. It shows measurement curves of a realizedoscillator that is using the present invention. The top curve displaysthe drain-source-voltage of the transistor 1 that is used in theoscillator circuit of FIG. 10. If the voltage is zero, the oscillator isturned off. If the voltage rises, the oscillator is switched-on. At timezero the Drain-Source-Voltage is turned on from 0V to 2.8V. At thispoint in time the oscillator starts its operation, i.e. the generationof an oscillation. The lower curve displays the DC-part of thetransistors input. This voltage is for gallium nitride HEMTs negative.In our example −4V. The inventive usage of a device inhibiting apositive current to reach the gate electrode 1 a, realized preferably bya diode in the transistors input network gives the circuit thepossibility to self-adjust its gate-DC-voltage. In FIG. 11, thegate-voltage changes at t=0 from −4 V to −7 V. This variation is notadded by the DC power supply 4. This reduction of the voltage level isdone by the oscillator circuit itself through self-biasing of the biaspoint of the field effect transistor 1. The data are observed passivelywith an oscilloscope. The Gate-Voltage changes to become more negativeby the oscillators signal itself. The variation in thegate-source-voltage shifts the bias point from class A to C by passingAB and B. A self-adjusting class of operation is the effect.

During switch-on, the oscillator microwave amplitude rises up at thetransistors gate 1 a. The large transistor input amplitude is generatinga positive DC current due to the effect of non-linear inputcharacteristic. It is an effect of partly rectification. That means thevoltage amplitude at the transistors internal gate-source-diode isgenerating an asymmetric current. Asymmetry matters: the current has aDC-fraction. This DC-current cannot flow continuously since there is noDC-path to anywhere, i.e. not to the feedback circuit and also not tothe gate-voltage supply 4 because of the invented usage of a diode 5.The diode 5 of the gate bias network prevents a fixed DC-gate-voltage.The temporary DC-current is recharging the bias-T's 6 filter capacitoruntil the current is zero. During recharging, the voltage is changedtowards negative. The speed of changing the voltage is given by theratio between the DC-current that is generated by the transistors inputcharacteristic and the value of bias-T's filter capacitor:

$\frac{U_{{{Gate} \cdot D}\; C}}{t} = {- \frac{I_{g,{D\; C}}}{C_{1}}}$

The recharging of the capacitor C1 and the following voltage variationhas a strong influence on the circuit. At first the bias point changes.For this reason also the efficiency, linearity, DC-supply and microwaveoutput power will change. The class of operation also changes.

In the given second embodiment, the bias point is shifted in a way thatchanges it class from A to C. A usage of the nice start-up condition ofclass A and shifting the bias point into class C region for betterefficiency. Oscillators in large signal regime is the main focus.

Thus, this invention offers new capabilities in designing oscillatorsand power amplifiers.

LIST OF REFERENCE SIGNS

-   1 field effect transistor    -   1 a gate electrode    -   1 b drain electrode    -   1 c source electrode-   2 load-   3 radio frequency network-   4 gate bias network-   5 device having non-linear characteristic-   6 gate bias-T-   7 drain bias-T-   8 drain bias network-   9 power amplifier-   10 feedback circuit/filter-   11 gate DC supply-   12 drain DC supply-   13 RF block-   14 DC block

1. A bias network for field effect transistors comprising: a fieldeffect transistor (1), the field effect transistor having an internalgate-diode, a source electrode (1 c) connected to ground and a drainelectrode (1 b) connected to a load (2); a radio frequency network (3)connected to the gate electrode (1 a) of the field effect transistor(1); a gate bias network (4) connected to the gate electrode (1 a) ofthe field effect transistor (1); characterized in that a device having anon-linear characteristic (5) is provided in series between the gateelectrode (1 a) of the field effect transistor (1) and the gate biasnetwork (4), the radio frequency network (3) and the gate bias network(4) are connected to the gate electrode (1 a) of the field effecttransistor (1) via a gate bias-T device (6) comprising a capacitor (C1)and wherein the device having non-linear characteristic (5) is providedbetween the gate bias-T device (6) and the gate bias network (4).
 2. Thebias network for field effect transistors according to claim 1, whereinthe gate bias network (4) provides a negative DC voltage to the gateelectrode (1 a) of the field effect transistor (1) and the device havinga non-linear characteristic (5) is arranged such that a negative currentpasses through the device (5) but a positive current is blocked orreduced.
 3. The bias network for field effect transistors according toclaim 1, wherein the device having a non-linear characteristic (5)comprises a diode or a transistor or a network having the characteristicof a diode.
 4. The bias network for field effect transistors accordingto claim 1, wherein the field effect transistor (1) is a HEMTtransistor.
 5. The bias network for field effect transistors accordingto claim 4, wherein the field effect transistor (1) is a GaN HEMTtransistor.
 6. The bias network for field effect transistors accordingto claim 1, wherein a second bias-T device (7) is provided between thedrain electrode (1 b) and the load (2) as well as between the drainelectrode (1 b) and a drain bias network (8).
 7. The bias network forfield effect transistors according to claim 1, wherein the radiofrequency network (3) inputs the output signal of a power amplifier (9)to the field effect transistor (1).
 8. The bias network for field effecttransistors according to claim 1, wherein the radio frequency network(3) inputs the output signal of a feedback network (10) to establish anoscillator network.
 9. The bias network for field effect transistorsaccording to claim 1, wherein the device (5) is realized as a diode anda zener diode is connected to a point between the diode (5) and the gatebias-T (6) such that the DC voltage is limited in the negativedirection.
 10. Method for self-adjustment of a bias point of a fieldeffect transistor in RF applications, comprising the steps of: providinga negative DC voltage by a gate bias network (4) via a gate bias-Tdevice (6) comprising a capacitor (C1) to a gate electrode (1 a) of afield effect transistor (1) having an internal gate diode, and providingan additional RF signal by a RF network (3) via the gate bias-T device(6) to the gate electrode (1 a) such that a negative current flowstowards the gate (1 b) of the field effect transistor (1); increasingthe RF signal of the RF network (3), whereupon the gate currentincreases to become positive; self-adjusting the bias point of thefield-effect transistor by blocking or limiting a positive current atthe gate electrode (1 a) using a device having non-linearcharacteristics (5) provided between the gate bias-T device (6) and thegate bias network (4).
 11. Method for self-adjustment of a bias point ofa field effect transistor in RF applications of claim 10, wherein anoscillator feedback-circuit is connected to the gate electrode (1 a) ofthe transistor (1).
 12. Method for self-adjustment of a bias point of afield effect transistor in RF applications of claim 10, furthercomprising the step of: limiting the maximum negative value of theself-adjustment of the bias point by providing a zener diode between thedevice with non-linear characteristics (5) and the gate electrode (1 a).13. (canceled)